"High efficiency linear regulators." 12 pages.
This app note discusses tricks to improve the efficiency of linear regulators by decreasing the input-to-output voltage drop. For example, Figure 5 shows a SCR preregulator for an AC-to-DC regulator, similar to Figure 7 in App Note 2. The SCR circuit keeps the voltage drop across the LT1086 around 2V, thus improving efficiency. (There are a lot of circuits here that are improved versions of circuits from App Note 2.)
Most of these applications involve using a DC-to-DC switching regulator in front of the linear regulator to control the voltage drop. We've also seen this approach previously (see App Note 2 Figure 5 and App Note 29 Figure 46). Figure 8a shows a DC-to-DC regulator scheme. Figure 8a has two switching regulators, one to control the voltage (6.75V) at the input of the LT1083, and another to provide 30V to overdrive the gate of the main switch. This application has a fixed output voltage, so the main switching regulator controls the input voltage of the LT1083. Figure 8b is used where the output is variable, and the feedback path of the main switching regulator measures the voltage drop across the LT1083.
Figure 11 is a linear regulator design with only 400-mV drop out. The switching regulator in this schematic provides a large voltage only for the gate of the pass transistor. The best circuit, shown in Figure 13, combines Figures 8a and 11 into a regulator with a linear output and an efficiency between 76% and 86%. The last circuit, shown in Figure 15, is a micropower version of the previous circuits, using the LT1020 linear regulator and using its integrated comparator to implement the switching regulator loop.
Appendix A, Achieving Low Dropout, discusses the tradeoffs involved in choosing the right device for the pass element in a linear regulator. The final sentence on page AN32-10 is a close runner-up for best quote: "Readers are invited to submit results obtained with our emeritus thermionic friends, shown out of respectful courtesy." (For an example, see Figure 11A in App Note 2.)
Appendix B discusses the LT1083 family of low drop out regulators. Appendix C discusses the measurement of power consumption and shows a circuit for measuring the instantaneous power in a 120V line. This circuit is a very useful instrumentation scheme. Unfortunately, the Analog Devices 286J isolation amplifier is no longer available (and the link to the datasheet on Analog Devices' website is broken). However, 286J amplifiers are still available from second sources, but I don't have any experience with them.
The best quote is actually a Freudian slip on pages AN32-8 and 9: "A drop at the pre-regulator's output (Pin 3 of the LT1020 regulator, Trace A, Figure 16) causes the LT1020's comparator to go high. The 74C04 inverter chain switches, biasing the P-channel MOSFET switch's grid (Trace B)." Of course, he meant "gate" instead of "grid", but I think we can forgive him for having vacuum tubes on the brain. (I sincerely hope that Linear Technology never fixes this typo!)
Showing posts with label Vacuum tubes. Show all posts
Showing posts with label Vacuum tubes. Show all posts
09 September 2011
19 August 2011
App Note 23
"Micropower circuits for signal conditioning." 24 pages.
This app note discusses techniques for designing extremely low-power circuits. As Jim says, "Although micropower ICs are available, the interconnection of these devices to form a functioning micropower circuit requires care."
The first two circuits are low-power temperature sensors (the first with a thermistor and the second with a thermocouple). The next two circuits are strain-gauge amplifiers, which use sampling techniques to achieve micropower performance, exploiting low duty cycle to give low power consumption. (Many more bridge circuits are coming up in App Note 43.) The next three circuits are temperature-monitoring applications for the LTC1040, LTC1041, and LTC1042 family of micropower comparator circuits. Figure 6 is a clever 4mA-to-20mA current-loop thermistor amplifier that uses the current loop signal as the power source.
Figure 9 is a micropower SAR analog-to-digital converter. Successive approximation is a good technique for low-power A-to-D, but unfortunately, now that SA registers (like the 74C905 used here) have been discontinued, it's harder to implement in discrete form. Figure 11 shows a micropower single-slope A-to-D converter, which only consumes 100 microamps (the recently discontinued 74C906 could easily be replaced by another low-offset CMOS switch). Figure 14 is a micropower sample and hold (SAR A-to-D converters require a S&H front end). This circuit cleverly uses the programming pin on the LT1006 to turn down the power consumption in the hold mode.
The best circuit is Figure 16, a micropower 10-kHz voltage-to-frequency converter. This circuit is also known as "The Zoo Circuit" and Jim wrote a chapter in his first book dedicated to it. Note the quote on page AN23-13: "A nice day at the San Francisco Zoo…, instrumental in arriving at the final configuration, is happily acknowledged." Rather than discuss the circuit here, I'll wait until I review his first book. (However, I will note here that there is an error in the schematic: the base of Q4 should be connected to its collector.) Figure 20 is a higher-speed V-to-F converter, reaching 1 MHz.
The final circuits are power regulators. Figure 22 is a switching regulator (using a discontinued 74C907 as the switch, we've got the whole family now). Figure 25 is a switching regulator that maintains a constant voltage drop across the LT1020 regulator, using the integral comparator. I like the start-up circuitry here. Figures 28 through 31 show off other tricks using the LT1020.
The box sections at the end (why not appendices?) cover a number of topics. Box Section A discusses low-power techniques and the design evolution of the zoo circuit. Box Section B discusses the LTC1040, LTC1041, and LTC1042 family of micropower comparator circuits. And finally, Box Section C discusses the effects of test equipment on micropower circuits (or, stated another way, suggestions that might power your circuits from the input source: just turn up the amplitude on that pulse generator).
Best quote (from page AN23-19): For example, everyone "knows" that "MOS devices draw no current." Unfortunately, Mother Nature dictates that as frequency and signal swings go up, the capacitances associated with MOS devices begin to require more power. It is often a mistake to automatically associate low power operation with a process technology. While it's likely that CMOS will provide lower power operation for a given function than 12AX7s, a bipolar approach may be even better.
This app note discusses techniques for designing extremely low-power circuits. As Jim says, "Although micropower ICs are available, the interconnection of these devices to form a functioning micropower circuit requires care."
The first two circuits are low-power temperature sensors (the first with a thermistor and the second with a thermocouple). The next two circuits are strain-gauge amplifiers, which use sampling techniques to achieve micropower performance, exploiting low duty cycle to give low power consumption. (Many more bridge circuits are coming up in App Note 43.) The next three circuits are temperature-monitoring applications for the LTC1040, LTC1041, and LTC1042 family of micropower comparator circuits. Figure 6 is a clever 4mA-to-20mA current-loop thermistor amplifier that uses the current loop signal as the power source.
Figure 9 is a micropower SAR analog-to-digital converter. Successive approximation is a good technique for low-power A-to-D, but unfortunately, now that SA registers (like the 74C905 used here) have been discontinued, it's harder to implement in discrete form. Figure 11 shows a micropower single-slope A-to-D converter, which only consumes 100 microamps (the recently discontinued 74C906 could easily be replaced by another low-offset CMOS switch). Figure 14 is a micropower sample and hold (SAR A-to-D converters require a S&H front end). This circuit cleverly uses the programming pin on the LT1006 to turn down the power consumption in the hold mode.
The best circuit is Figure 16, a micropower 10-kHz voltage-to-frequency converter. This circuit is also known as "The Zoo Circuit" and Jim wrote a chapter in his first book dedicated to it. Note the quote on page AN23-13: "A nice day at the San Francisco Zoo…, instrumental in arriving at the final configuration, is happily acknowledged." Rather than discuss the circuit here, I'll wait until I review his first book. (However, I will note here that there is an error in the schematic: the base of Q4 should be connected to its collector.) Figure 20 is a higher-speed V-to-F converter, reaching 1 MHz.
The final circuits are power regulators. Figure 22 is a switching regulator (using a discontinued 74C907 as the switch, we've got the whole family now). Figure 25 is a switching regulator that maintains a constant voltage drop across the LT1020 regulator, using the integral comparator. I like the start-up circuitry here. Figures 28 through 31 show off other tricks using the LT1020.
The box sections at the end (why not appendices?) cover a number of topics. Box Section A discusses low-power techniques and the design evolution of the zoo circuit. Box Section B discusses the LTC1040, LTC1041, and LTC1042 family of micropower comparator circuits. And finally, Box Section C discusses the effects of test equipment on micropower circuits (or, stated another way, suggestions that might power your circuits from the input source: just turn up the amplitude on that pulse generator).
Best quote (from page AN23-19): For example, everyone "knows" that "MOS devices draw no current." Unfortunately, Mother Nature dictates that as frequency and signal swings go up, the capacitances associated with MOS devices begin to require more power. It is often a mistake to automatically associate low power operation with a process technology. While it's likely that CMOS will provide lower power operation for a given function than 12AX7s, a bipolar approach may be even better.
08 August 2011
App Note 18 part 1
"Power gain stages for monolithic amplifiers." 16 pages.
This app note contains several different discrete output stages for op amps. The three major themes here are high output current, high output (rail-to-rail) swing, and high output voltage.
The first major theme is current boosting. The first circuits (in Figure 1) exploit Widlar's LT1010 power buffer (up to 150 mA), in the application for which it was designed. I really like the circuit in Figure 2, which uses the power-supply terminals for unintended purposes. By sensing the current in the supply pins, you can tell whether the output buffer is sourcing or sinking current, and then drive huge currents (up to 3 amps in the MJE2955 and MJE3055) based on that measurement. The power-supply pins don't always have to connected to just power! This circuit is a nice reminder than buffers are really four-terminal devices (and op amps are five-terminal devices), and all of the terminals can be used in a clever design.
Figure 3 is a fast output stage, using a feed-forward path, similar to Figure 8 in App Note 6. The op amp is being used as a low-frequency error servo, while the feed-forward through the JFET provides the high-frequency path, with a slew rate of one thousand volts per microsecond.
The second major theme is "voltage-gain" stages for nearly rail-to-rail output swing. Using CMOS inverters as "linear" gain elements (as in Figure 5a) weirds me out. I just can't get over my distrust of digital circuits to use them this way. Is the gate behavior in the linear region reliable enough? I guess so. Figure 5b uses bipolar transistors to drive closer to the rails at higher currents. The circuits in Figure 5 are run off a five-volt rail; Figure 7 is another (nearly) rail-to-rail output stage, this time for plus-and-minus 15V rails.
The third major theme is high-voltage output stages, with four example circuits. Figure 9 is roughly similar to some of the other output buffers, but using high-voltage transistors and driving the output node to plus-and-minus 125 volts. (I appreciate the comment that the input common-mode voltage limits require a minimum gain of 11 in the non-inverted connection. In other words, "Remember to do the math!")
Figure 11 is a high-voltage stage, similar to Figure 9, but that uses vacuum tubes. (There aren't many (modern) op-amp circuits that require a 12.6VAC filament supply.) Unfortunately, he calls them "Mr. De Forest's Descendants". I know that he is trying to funny, but De Forest deserves no credit for the invention of the vacuum tube. Don't get me started (instead, I'll just refer you to to Chapter 1 of "The Design of CMOS Radio-Frequency Integrated Circuits" by Thomas Lee).
Figure 13 is an extremely high-voltage output stage, driving up to +1000 volts, but powered just from +28V. The basic trick here is the integral boost switching regulator and the transformer. The current limiting is done by the comparator C1 and the diode network, which brute-forces off the oscillator, the darlington drive, and the drives to the MOSFETs. Finally, Figure 15 is implements a bipolar high-voltage step-up stage, by restoring the "polarity" of the output voltage after the transformer and rectifier with a SCR-based synchronous demodulator.
Best quote (from page AN18-8): "The transistor inverter [in Figure 11] is necessitated because our thermionic friends have no equivalent to PNP transistors."
I'll discuss the box section on frequency compensation next time.
Related:
This app note contains several different discrete output stages for op amps. The three major themes here are high output current, high output (rail-to-rail) swing, and high output voltage.
The first major theme is current boosting. The first circuits (in Figure 1) exploit Widlar's LT1010 power buffer (up to 150 mA), in the application for which it was designed. I really like the circuit in Figure 2, which uses the power-supply terminals for unintended purposes. By sensing the current in the supply pins, you can tell whether the output buffer is sourcing or sinking current, and then drive huge currents (up to 3 amps in the MJE2955 and MJE3055) based on that measurement. The power-supply pins don't always have to connected to just power! This circuit is a nice reminder than buffers are really four-terminal devices (and op amps are five-terminal devices), and all of the terminals can be used in a clever design.
Figure 3 is a fast output stage, using a feed-forward path, similar to Figure 8 in App Note 6. The op amp is being used as a low-frequency error servo, while the feed-forward through the JFET provides the high-frequency path, with a slew rate of one thousand volts per microsecond.
The second major theme is "voltage-gain" stages for nearly rail-to-rail output swing. Using CMOS inverters as "linear" gain elements (as in Figure 5a) weirds me out. I just can't get over my distrust of digital circuits to use them this way. Is the gate behavior in the linear region reliable enough? I guess so. Figure 5b uses bipolar transistors to drive closer to the rails at higher currents. The circuits in Figure 5 are run off a five-volt rail; Figure 7 is another (nearly) rail-to-rail output stage, this time for plus-and-minus 15V rails.
The third major theme is high-voltage output stages, with four example circuits. Figure 9 is roughly similar to some of the other output buffers, but using high-voltage transistors and driving the output node to plus-and-minus 125 volts. (I appreciate the comment that the input common-mode voltage limits require a minimum gain of 11 in the non-inverted connection. In other words, "Remember to do the math!")
Figure 11 is a high-voltage stage, similar to Figure 9, but that uses vacuum tubes. (There aren't many (modern) op-amp circuits that require a 12.6VAC filament supply.) Unfortunately, he calls them "Mr. De Forest's Descendants". I know that he is trying to funny, but De Forest deserves no credit for the invention of the vacuum tube. Don't get me started (instead, I'll just refer you to to Chapter 1 of "The Design of CMOS Radio-Frequency Integrated Circuits" by Thomas Lee).
Figure 13 is an extremely high-voltage output stage, driving up to +1000 volts, but powered just from +28V. The basic trick here is the integral boost switching regulator and the transformer. The current limiting is done by the comparator C1 and the diode network, which brute-forces off the oscillator, the darlington drive, and the drives to the MOSFETs. Finally, Figure 15 is implements a bipolar high-voltage step-up stage, by restoring the "polarity" of the output voltage after the transformer and rectifier with a SCR-based synchronous demodulator.
Best quote (from page AN18-8): "The transistor inverter [in Figure 11] is necessitated because our thermionic friends have no equivalent to PNP transistors."
I'll discuss the box section on frequency compensation next time.
Related:
12 July 2011
App Note 2
Ha! Another classic element of the Jim Williams style: Along with more scope photos from his Tek 547, this app note includes a vacuum-tube circuit in Figure 11A. Plus, it's an Eimac 75TH (which is a gigantic bulbous triode, almost eight inches tall; not really a practical tube to use). This is Jim's practical-joker side.
This app note contains more tricks and tips for using three-terminal regulators. Figure 2 (particularly the presence of the big capacitors) supports my assertion that feedback-loop design with three-terminal regulators is hard and ad hoc because good transfer-function models aren't available. In this case, you've got to add a lot of damping to the loop to make it well behaved. Then, once you've got 100 microfarads on the output, you've got to add Q4 to maintain a quick disable. Similarly, Figure 12, showing an LT1001 precision op amp in the feedback path of an LT317AH, gives me some more feedback-induced indigestion. What's the loop gain around that loop? Yikes.
Figure 5 uses a switching regulator to control the voltage across a linear regulator. Neat trick! The best circuit is the switcher-controlled linear regulator in Figure 7, driven from AC using SCRs that is up to 85% efficient. (I must admit that I have never used an SCR in a circuit design. I am embarrassed.) Note the LM301A in an integrator topology with 100-pF compensation cap and a diode clamp on pin 8. Again, the loop compensation here is very, very conservative.
The final circuit is another 110VAC/220VAC dual voltage solution (a much better approach than Figure 5 in App Note 1). Clearly, he was grappling with this problem. Plus, another SCR!
Best quote (page AN2-5): "Because these two feedback systems are interlocked, frequency compensation can be difficult." No argument here.
UPDATE (with respect to Figure 12, and the LT1001 and LT317A): As Joe alludes to in his comment below, there are basically two ways to build an adjustable regulator, as illustrated here (with much simplification).
UPDATE (with respect to Figure 12, and the LT1001 and LT317A): As Joe alludes to in his comment below, there are basically two ways to build an adjustable regulator, as illustrated here (with much simplification).
The topology on the left would be totally unstable with an op amp in the feedback path, while the topology on the right might be stable with an op amp in the feedback path (depending on the unity-gain frequency of the op amp and the bandwidth of the regulator). I was thinking that the LT317A was like the topology on the left, but looking at the datasheet, it's like the topology on the right (either way, it'd be nice to have a transfer-function model). Thanks Joe!
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